1. Field of the Invention
The present invention relates to an inverter controller which controls inverters having self-turn-off switching devices.
2. Description of the Related Art
The performance of speed controllers for alternating-current motors depends largely on their current control efficiency. For high performance, the output current of the inverter must respond rapidly to the value of the current command sent from the external speed control system, torque control system, or other device.
In pulse width modulation (PWM) inverters, such rapid current response can be obtained using a PWM control method using the current hysteresis band. The control configuration in such a case is shown in FIG. 1.
In FIG. 1, 1 is a direct current power supply, 2 is a capacitor, and 3 is an inverter consisting of self-turn-off switching devices SUP, SVP, SWP, SUN, SVN, and SWN and respective diodes DUP, DVP, DWP, DUN, DVN, and DWN, to which they are connected in inverse parallel. 4 is an alternating-current motor. 5U, 5V, and 5W are Hall CTs. 6 is a current detector. 7U, 7V, and 7W are subtracters which output the respective deviations .DELTA.iu, .DELTA.iv, and .DELTA.iw on receiving respective phase current command values iu*, iv*, and iw* which should flow through the alternating-current motor and the alternating-current motor phase current detection values iu, iv, and iw output from the current detector. 8U, 8V, and 8W are hysteresis comparators which, on receiving current deviations .DELTA.iu, .DELTA.iv, and .DELTA.iw, change their output to logical value 1 if the deviation exceeds the set hysteresis width (hys/2) and change their output to logical value 0 if the deviation is less than (-hys/2). The outputs of hysteresis comparators 8U, 8V, and 8W are PWM signals Uo, Vo, and Wo of each of the phases of inverter 3 and are output to logic circuit 9. PWM signals Uo,Vo, and Wo are input into logic circuit 9, which then performs the prescribed on-delay-time processing of these signals and of their logical inverse signals then outputs drive signals to the six self-turn-off switching devices which make up inverter 3. The output from logic circuit 9 is sent via gate circuit 10 to the gates of the self-turn-off switching devices corresponding to inverter 3.
The operation of this type of circuit can be explained simply with reference to FIG. 2. Either side of the sine-wave current command value iu*, shown as a chain line, are the two hys/2 hysteresis widths, shown as dotted lines. If current iu changes to the positive direction and reaches the current command value (iu*+hys/2), the hysteresis comparator output becomes logical value 0 and the inverter negative-side switch comes on, causing negative voltage -Ed/2 to be applied to the motor coil and current iu to change to the negative direction, where Ed is the voltage of direct-current power supply 1. If, in changing to the negative direction, current iu reaches the current command value (iu*-hys/2), the positive-side switch comes on, causing positive voltage +Ed/2 to be applied to the motor coil and current iu to change to the positive direction. In this way, current iu can be controlled so that it is maintained within hysteresis width .+-.hys/2 of current reference iu*. Since the comparator gain may be considered almost infinite, very rapid current control response is possible. However, by the same token, a hysteresis width to limit the switching frequency is required. The highest switching frequency is determined by the leakage impedance of the coil, the direct current voltage of the inverter, the switching speed and other characteristics of the arc-self-suppressing switching elements, and the hysteresis width.
The above explanation is valid if one phase only is considered, and it is not possible to obtain an ideal PWM waveform as shown in FIG. 2 in an alternating-current motor drive using a three-phase inverter.
FIG. 3 shows some examples of waveforms when hysteresis-band PWM control is used in an alternating-current motor drive using a three-phase inverter. The waveforms of currents iu, iv, and iw are shown with their hysteresis bands along with the waveforms of motor phase voltages Vu, Vv, and Vw and the line voltage Vu-v. It is clear from the figure that the switching frequency changes. The switching frequency is low during period T1 and high during period T2. This kind of phenomenon occurs at low rotation speeds and when the current command value is small.
The low switching frequency during T1 is due to the fact that, since the voltage is +Ed/2 in all three phases, and all of the line voltages are 0, there is a period during which the current changes as a result of back electromotive force only, and since the rotation speed is low and therefore the back electromotive force is small, the current changes gradually. Furthermore, since the voltages of the three phases have the same potential and the magnitude of the vector resulting from the combination of these voltages such that all line voltages become zero is zero, the vector is hereinafter called the zero-voltage vector.
Conversely, the high switching frequency during T2 is due to the fact that the current changes rapidly, since at no time are the voltages of the three phases equal, and any two of motor line voltages Vu-v, Vv-w, and Vw-u are not zero. The ratio of the current ripple to the switching frequency during T2 is much higher than during T1. Voltage vectors which are not zero-voltage vectors are hereinafter called nonzero-voltage vectors.
High switching frequencies, as during T2, are more likely when the current reference (which is also called the current command in some cases hereinafter) is small, while low switching frequencies, as during T1, are more likely at low motor rotation speeds, when the back electromotive force is small. A low rotation speed combined with a small current reference will cause very large modulation frequency fluctuations.
Despite the advantage of very rapid current control response, hysteresis-band PWM control has recently fallen out of favor for the following main reasons:
(1) The switching frequency varies greatly, as during T1 and T2 in FIG. 3.
(2) During T2 in FIG. 3, at the same switching frequency the current ripple is considerably larger than that in triangle-wave-comparison PWM control and other types of PWM control.
It is difficult to counter ambient noise and radio noise if the modulation frequency varies. Furthermore, because the inverter must be capable of accommodating the highest modulation frequency, fluctuations in modulation frequency mean larger equipment. Widening the hysteresis band to lower the highest modulation frequency increases the magnitude of the current ripple. Hysteresis-band PWM control already has a larger current ripple than triangle-wave-comparison PWM control and other types of PWM control and therefore has the serious drawback of having to provide a large margin in the current ratings of the switching elements. As shown in FIG. 3, the switching frequency varies even though the rotation speed, the amplitude, and the hysteresis band are all fixed.
A method of solving these problems was published in `A current-control PWM inverter capable of harmonic suppression and rapid current response` by Satoshi Ogasawara, Tomoaki Nishimura, Hirobumi Akamoto, and Akira Nanba (Electrical Society Paper B, February 1986). If the current deviation is large, as in stepped current reference variation, the method uses conventional hysteresis-band PWM control. If the current deviation is small, as in a steady state, the method changes over to switching capable of harmonic suppression.
The steady-state method controls the switching by detecting the angle of the back electromotive force vector of the alternating-current motor and using only the two nonzero-voltage vectors either side of the back electromotive force vector (those with the nearest angles) and the zero-voltage vector. It is believed that harmonic suppression is more effective if voltage vectors with a small current variation ratio are selected. However, as the motor rotation speed increases, the voltage drop due to inductance increases, and, even in a steady state, a voltage vector whose angle differs greatly from that of the back electromotive force vector must be used to obtain the desired current. In the present invention, when the current deviation becomes large, the method changes over to conventional hysteresis-band PWM control.
A method using deviation current differentiation and, for permanent-magnet motors, a method using calculation of the back electromotive force are presented as methods of detecting the back electromotive force vector. However, as described in the above publication, the method using deviation current differentiation is sensitive to noise, while the method of calculating the back electromotive force for permanent-magnet motors assumes a steady state, leaving open the possibility of an error in the transition state. Because this switching method determines the switching signal on the basis of the direction of the back electromotive force vector, erroneous detection of the back electromotive force vector directly results in the output of an incorrect switching signal.
The above method must therefore be used with conventional hysteresis-band PWM control. Even if rapid response is not required, it is not sufficient to use only a switching method capable of harmonic suppression.
All the same, because of the large harmonic waves and the very serious problem of modulation frequency fluctuations, the use of hysteresis-band PWM control has become confined to special applications. Also, in the method based on the detection of the back electromotive force vector, the detection of the back electromotive force itself constitutes a problem. Where the back electromotive force load is unknown, the back electromotive force is estimated using deviation current differentiation, resulting in a system that is vulnerable to noise.